Rectifying/smoothing circuit and double-ended converter

ABSTRACT

A rectifying and smoothing circuit ( 3 ) which includes a first inductor connected between one end ( 2   b   1 ) of an output winding of a transformer ( 2 ) and a low potential-side output portion ( 16   b ), a second inductor connected between the other end ( 2   b   2 ) of the output winding and the low potential-side output portion, a first rectifying element ( 11 ) connected between the one end of the output winding and a high potential-side output portion ( 16   a ), and a second rectifying element ( 12 ) connected between the other end of the output winding and the high potential-side output portion, and which generates a DC voltage (V0) by rectifying and smoothing a bipolar voltage (VS) induced across the output winding, has a construction permitting reduction of the size and manufacturing costs thereof and enhancement of smoothing effects. In this rectifying and smoothing circuit, the first and second inductors are constructed by a first winding ( 17   a ) and a second winding ( 17 ) of a transformer, which are wound in a manner permitting magnetic fluxes generated by respective currents flowing therethrough to cancel each other.

TECHNICAL FIELD

This invention relates to a rectifying and smoothing circuit based on acurrent doubler rectification method, and a double-ended (bipolar)converter using the rectifying and smoothing circuit, such as apush-pull converter, a half-bridge converter, an asymmetric half-bridgeconverter, a full-bridge converter, and an active clamp converter.

BACKGROUND ART

Conventionally, as a power supply including a rectifying and smoothingcircuit based on the current doubler rectification method of theabove-mentioned kind, a power supply 81 shown in FIG. 17 is known. Thispower supply 81 includes a switching transformer 2, and a currentdoubler rectifying and smoothing circuit 82. In this case, the currentdoubler rectifying and smoothing circuit 82 is comprised of a smoothingchoke coil 14 connected between one end 2 b 1 of a secondary winding 2 bof the transformer 2 and an output terminal 16 b on a low-potentialside, a smoothing choke coil 15 connected between the other end 2 b 2 ofthe secondary winding 2 b and an output terminal 16 b on ahigh-potential side and having the same inductance value as that of thechoke coil 14, a diode 11 as a rectifying element, connected between theone end 2 b 1 of the secondary winding 2 b and an output terminal 16 a,and a diode 12 as a rectifying element, connected between the other end2 b 2 of the secondary winding 2 b and the output terminal 16 a. Thecurrent doubler rectifying and smoothing circuit 82 outputs a DC voltageV0 generated by rectifying and smoothing a bipolar voltage inducedbetween the opposite ends of the secondary winding 2 b to a load 4.

In this power supply 81, push-pull FET circuits, not shown, connected toone end 2 a 1 of a primary winding 2 a of the transformer 2 and theother end 2 a 2 thereof, respectively, are driven at 180 degrees out ofphase with respect to each other, whereby as shown in FIG. 18, a bipolarvoltage VS having a voltage value±VS is induced between the oppositeends of the secondary winding 2 b of the transformer 2. In this case, ina period T1 during which one of the FET circuits is controlled to an ONstate at a duty ratio D of 25%, a high voltage is induced on the side ofthe one end 2 b 1 of the secondary winding 2 b during the ON time periodTON of the FET, and this induced voltage causes a current I31 shown inFIG. 17 to flow through a current path of the one end 2 b 1 of thesecondary winding 2 b, the diode 11, the load 4, the choke coil 15, andthe other end 2 b 2 of the secondary winding 2 b. In this state, asshown in FIG. 18, a voltage VL15 having a voltage value(VS−V0=(1−D)/D−V0/f, where f represents a frequency of the bipolarvoltage VS) and directed as shown in FIG. 17 is generated betweenopposite ends of the choke coil 15, whereby energy is accumulated in thechoke coil 15.

Further, during an OFF time period TOFF of the period T1, the energyaccumulated in the choke coil 15 causes a current I32 to flow in adirection shown in the same figure through a current path of one end ofthe choke coil 15, the diode 12, the load 4, and the other end of thechoke coil 15. Consequently, the voltage VL15 between the opposite endsof the choke coil 15 is caused to have a voltage (−V0), and at the sametime, as shown in FIG. 18, a current IL15 varying within a range of acurrent variation width ((VS−V0)•TON/Lo=(1−D)•V0/f, where Lo representsan inductance value of the choke coils 14 and 15) flows through thechoke coil 15.

Further, in the period T2 (the same time period as the period T1) duringwhich the other FET is controlled to an ON state at a duty ratio D of25%, a high voltage is induced on the side of the other end 2 b 2 of thesecondary winding 2 b during the ON time period TON of the FET, and thisinduced voltage causes a current I33 shown in FIG. 17 to flow through acurrent path of the other end 2 b 2 of the secondary winding 2 b, thediode 12, the load 4, the choke coil 14, and the one end 2 b 1 of thesecondary winding 2 b. In this state, as shown in FIG. 18, between theopposite ends of the choke coil 14 is generated a voltage VL14 having avoltage value (VS−V0) and directed as shown in FIG. 17, whereby energyis accumulated in the choke coil 14.

Further, during an OFF time period TOFF of the period T2, the energyaccumulated in the choke coil 14 causes a current I34 to flow in adirection shown in FIG. 17 through a current path of one end of thechoke coil 14, the diode 11, the load 4, and the other end of the chokecoil 14. Consequently, the voltage VL14 between the opposite ends of thechoke coil 14 become equal to a voltage value (−V0), and as shown inFIG. 18, a current IL14 varying within a range of a current variationwidth ((VS−V0)•TON/Lo=(1−D)•V0/f) flows through the choke coil 14. Inthe above process of operation, each of average current values of thecurrents IL15 and IL14 becomes equal to one half of an output currentI0, since a sum total of the current values of the currents becomesequal to the output current I0, shown in FIGS. 17 and 18, and at thesame time the current values thereof are equal to each other. It shouldbe noted that as shown in FIGS. 17 and 18, a ripple current IC flowingthrough the capacitor 13 varies within a range of a current variationwidth ((1−2D)•V0/f=(1−TON/(T−TON))•(VS−V0)•TON/Lo, where D represents aduty ratio, and f represents the reciprocal of the period T).

As described above, smoothing operations are carried out by the chokecoils 14 and 15 during a time period of each of the periods T1 and T2,so that as shown in FIG. 18, an output current Io from which a ripplecomponent is substantially eliminated is output to the load 4.

DISCLOSURE OF THE INVENTION

The inventor studies the above prior art and found out the followingproblems:

Firstly, the conventional current doubler rectifying and smoothingcircuit 82 uses the choke coils 14 and 15 constructed as separatecomponent parts independent of each other. Therefore, the currentdoubler rectifying and smoothing circuit 82 has a large number ofcomponent parts and suffers from the problem of increased manufacturingcosts caused by the mounting of component parts.

Secondly, a DC current having a predetermined current value constantlyflows through the choke coils 14 and 15, as shown in FIG. 18,respectively. In this case, as shown by the characteristics of the DCcurrent with respect to the excitation inductance (LX) in FIG. 16, thereis a predetermined relationship between the excitation inductance (LX)of smoothing coils (choke coils 14 and 15 in the above current doublerrectifying and smoothing circuit 82) and a DC current allowed to passthrough the smoothing coils. That is, to enhance the effect of thesmoothing coils as smoothing filters, it is preferred that the smoothingcoils have a large excitation inductance, whereas the value of asaturation DC current is reduced as the excitation inductance isincreased. More specifically, when the effective volume of a smoothingchoke coil as a magnetic material is small, if the excitation inductanceis set to a small value (L2), as shown by a characteristic CH1, thesmoothing choke coil can be used without magnetic saturation thereofuntil a current having a rather large current value (I22) flowstherethrough, whereas if the excitation inductance is set to a largevalue (L1), there is a fear that the magnetic material undergoesmagnetic saturation, since the limit value of a current below which thesmoothing coil can be used without magnetic saturation thereof islowered to a very small value (I21). Further, as shown by acharacteristic CH2, when the effective volume of the coil as a magneticmaterial is made sufficiently large, if the excitation inductance is setto a small value (L2), the smoothing choke coil can be used withoutmagnetic saturation thereof, until a current having a rather largecurrent value (I24) flows therethrough, and furthermore, even if theexcitation inductance is set to a rather large value (L1), the smoothingchoke coil can be used without magnetic saturation thereof, until thecurrent flowing reaches a larger current value (I23) than in the case ofthe characteristic CH1. In such a case, however, a large mounting spaceis required.

For the above reason, the conventional current doubler rectifying andsmoothing circuit 82 can suffer from magnetic saturation of the chokecoils 14 and 15. In such a case, the choke coils 14 and 15 and thepush-pull FET circuits on the primary side can be broken or the chokecoils 14 and 15 cease to function as smoothing filters. On the otherhand, to cause the choke coils 14 and 15 to fully perform the functionas smoothing filters without causing magnetic saturation thereof, it isrequired that the effective volumes of the coils as magnetic materialsare made sufficiently large. This results in an increase in the size ofthe current doubler rectifying and smoothing circuit 82, and further inan increase in the size of the power supply 81.

The present invention has been made to solve the above problems, and itis a main object of the invention to provide a rectifying and smoothingcircuit which can be reduced in size, and at the same time permitreduction of manufacturing costs through decreasing the number ofcomponent parts thereof and enhance a smoothing effect thereof, as wellas a double-ended converter using the rectifying and smoothing circuit.

The rectifying and smoothing circuit according to this inventionincludes a first inductor connected between one end of an output windingof a first switching transformer and a low potential-side outputportion, a second inductor connected between another end of the outputwinding and the low potential-side output portion, a first rectifyingelement connected between the one end of the output winding and a highpotential-side output portion, and a second rectifying element connectedbetween the another end of the output winding and the highpotential-side output portion, the rectifying and smoothing circuitgenerating a DC voltage by rectifying and smoothing a bipolar voltageinduced across the output winding, and is characterized in that thefirst inductor and the second inductor are constructed by a firstwinding and a second winding of a second transformer, respectively, thefirst winding and the second winding being wound in a manner permittingmagnetic fluxes generated by respective currents flowing therethrough tocancel each other.

In this rectifying and smoothing circuit, the first winding and thesecond winding of the second transformer serve as smoothing coils.Therefore, during the smoothing operation, magnetic fluxes generatedfrom the windings by respective currents flowing therethrough canceleach other, so that a DC component of the exciting current flowingthrough the second transformer becomes approximately equal to 0 A.Consequently, magnetic saturation in the second transformer caused by aDC bias can be prevented. This makes it possible to construct smoothingcoils having a large excitation inductance by using ferrite cores havingsmall effective volumes, and hence the current doubler rectifying andsmoothing circuit can be caused to serve as more excellent smoothingfilters. Further, since the two choke coils 14 and 15 of theconventional current doubler rectifying and smoothing circuit 82 arereplaced by one second transformer, the rectifying and smoothing circuitand further the power supply using this rectifying and smoothing circuitcan be made smaller in size.

In this case, the first inductance and the second inductor are eachconstructed by a series circuit of an equivalent leakage inductance ofthe second transformer and an equivalent excitation inductance thereof.

Further, it is preferred that the second transformer is constructed bythe first winding and the second winding wound in a manner spaced fromeach other by a predetermined distance. This construction of the secondtransformer permits the leakage inductance of the second transformer tobe defined to be equal to a desired value according to the separationdistance. Consequently, the effects of the second transformer assmoothing filters can be determined as desired depending on an object oruse of the transformer. Further, the second transformer may be formedwith a bypass passage for a magnetic flux. In this case as well, it ispossible to define the leakage inductance of the second transformer tobe equal to a desired value according to an amount of amagnetic fluxpassing through the bypass passage. Further, the second transformer mayuse magnetic cores having a low magnetic permeability. In this case aswell, the leakage inductance of the second transformer can be defined tobe equal to a desired value according to the degree of the magneticpermeability.

On the other hand, the second transformer can be constructed by windingthe first winding and the second winding around magnetic cores formedwith gaps. In this rectifying and smoothing circuit, the excitationinductance of the second transformer can be defined to be equal to adesired value according to the widths of the gaps, so that the effectsof the second transformer as smoothing filters can be determined asdesired depending on an object or use of the transformer.

Further, a third inductor may be connected in series with at least oneof the first winding and the second winding of the second transformer.In this rectifying and smoothing circuit, if the inductance value of thethird inductor is properly defined, the leakage inductance of the secondtransformer can be adjusted to a desired value with ease.

Additionally, it is preferred that a double-ended converter isconstructed by incorporating the above rectifying and smoothing circuit.This construction of the double-ended converter realizes a power supplyhaving more excellent smoothing filters and moreover is reduced inmanufacturing costs and size thereof by reduction of the number ofcomponent parts of the rectifying and smoothing circuit.

BRIEF DESCRIPTION OF DRAWINGS

FIG. 1 is a circuit diagram of a power supply 1 according to a firstembodiment;

FIG. 2 is a circuit diagram of a transformer 17 in the power supply 1according to the first embodiment;

FIG. 3 is a diagram showing an equivalent circuit of the transformer 17in FIG. 2;

FIG. 4 is a diagram showing an equivalent circuit of the transformer 17in FIGS. 2 and 3;

FIG. 5 is a diagram showing a voltage or current waveform of each ofcomponent parts of the power supply 1 according to the first embodiment,which includes a diagram showing a voltage waveform of a bipolar voltageVS induced between opposite ends of a secondary winding 2 b, a diagramshowing a voltage waveform of a voltage VLX2 between opposite ends of anexcitation inductance LX2, a diagram showing a voltage waveform of avoltage VLX1 between opposite ends of an excitation inductance LX1, adiagram showing a voltage waveform of a voltage (VLX1+VLL) betweenopposite ends of a first winding 17 a, a voltage waveform of a voltageVLL between opposite ends of a leakage inductance LL, a diagram showinga current waveform of a current Ib flowing through a second winding 17b, a diagram showing a current waveform of a current Ia flowing throughthe first winding 17 a, a diagram showing a current waveform of a ripplecurrent IC flowing through a capacitor 13, and a diagram showing acurrent waveform of an exciting current IT flowing through thetransformer 17;

FIG. 6A is a perspective view of an appearance of ferrite cores 22 and22 for use with the FIG. 2 transformer 17, and FIG. 6B is a perspectiveview of an appearance of the transformer 17;

FIG. 7A is a perspective view of an appearance of another transformer25, and FIG. 7B is a perspective view of an appearance of still anothertransformer 27;

FIG. 8 is a perspective view of an appearance of the transformer 17 witha coil 29 connected in series with a conducting wire 23 thereof;

FIG. 9 is a circuit diagram of a power supply 31 according to a secondembodiment, which is accompanied by a diagram showing current waveformsof switching signals SS1 and SS2 input to the power supply 31;

FIG. 10 is a diagram showing a voltage or current waveform of each ofcomponent parts of the power supply 31, which includes a diagram showinga voltage waveform of a bipolar voltage VS induced between opposite endsof a secondary winding 2 b, a diagram showing a voltage waveform of avoltage VLX2 between opposite ends of an excitation inductance LX2, adiagram showing a voltage waveform of a voltage VLX1 between oppositeends of an excitation inductance LX1, a diagram showing a voltagewaveform of a voltage (VLX1+VLL) between opposite ends of a firstwinding 17 a, a voltage waveform of a voltage VLL between opposite endsof a leakage inductance LL, a diagram showing a current waveform of acurrent Ib flowing through a second winding 17 b, a diagram showing acurrent waveform of a current Ia flowing through the first winding 17 a,a diagram showing a current waveform of a ripple current IC flowingthrough a capacitor 13, and a diagram showing a current waveform of anexciting current IT flowing through the transformer 17;

FIG. 11 is a circuit diagram of a power supply 41 according to a thirdembodiment;

FIG. 12 is a diagram showing current waveforms of switching signals SS3and SS4 input to the power supply 41 or the like;

FIG. 13 is a circuit diagram of a power supply 51 according to a fourthembodiment;

FIG. 14 is a circuit diagram of a power supply 61 according to a fifthembodiment;

FIG. 15 is a circuit diagram of a power supply 71 according to a sixthembodiment;

FIG. 16 is a characteristics diagram showing the relationship between aDC current and an excitation inductance LX;

FIG. 17 is a circuit diagram of a conventional power supply 81; and

FIG. 18 is a diagram showing a voltage or current waveform of each ofcomponent parts of the conventional power supply 81, which includes adiagram showing a voltage waveform of a bipolar voltage VS inducedbetween opposite ends of a secondary winding 2 b, a diagram showing avoltage waveform of a voltage VL15 between opposite ends of a choke coil15, a diagram showing a voltage waveform of a voltage VL14 betweenopposite ends of a choke coil 14, a diagram showing a current waveformof a current IL15 flowing through the choke coil 15, a diagram showing acurrent waveform of a current IL14 flowing through the choke coil 14, adiagram showing a current waveform of an output current I0, and adiagram showing a current waveform of a ripple current IC flowingthrough the capacitor 13.

BEST MODE OF CARRYING OUT THE INVENTION

A rectifying and smoothing circuit and a double-ended converteraccording to preferred embodiments of the invention will be describedbelow with reference to accompanying drawings.

(First Embodiment)

First, the operating principle of the rectifying and smoothing circuitaccording to the invention will be described with reference to FIG. 1.As shown in the figure, a power supply 1 includes a switchingtransformer 2 having a primary winding 2 a and a secondary winding 2 b,and a current doubler rectifying and smoothing circuit 3. The currentdoubler rectifying and smoothing circuit 3 is comprised of a diode 11 asa rectifying element connected between one end 2 b 1 of the secondarywinding 2 b and an output terminal 16 a, a diode 12 as a rectifyingelement connected between the other end 2 b 2 of the secondary winding 2b and the output terminal 16 a, and a capacitor 3 connected between theoutput terminals 16 a and 16 b. The current doubler rectifying andsmoothing circuit 3 outputs a DC voltage V0 generated by rectifying andsmoothing a bipolar voltage induced between opposite ends of thesecondary winding 2 b, to a load 4. Further, the current doublerrectifying and smoothing circuit 3 includes a transformer 17 which is aleakage transformer and corresponds to a second transformer of theinvention. In this embodiment, as shown in FIG. 2, the transformer 17actually includes a first winding 17 a having a number n1 of turns and asecond winding 17 b having a number n2 of turns. The windings 17 a and17 b are wound such that magnetic fluxes generated by respectivecurrents flowing therethrough cancel each other. Referring to FIG. 3,equivalently, the first winding 17 a of the transformer 17 isrepresented by a series circuit of a leakage inductance LL1 calculatedon the side of the first winding 17 a and an excitation inductance LX1calculated on the side of the first winding 17 a, and the second winding17 b of the transformer 17 is represented by a series circuit of aleakage inductance LL2 calculated on the side of the second winding 17 band an excitation inductance LX2 calculated on the side of the secondwinding 17 b.

The transformer 17 shown in FIG. 3 can be represented as an equivalentcircuit shown in FIG. 4. More specifically, in this transformer 17, theleakage inductance LL1 and the leakage inductance LL2 are represented asa leakage inductance LL calculated on the side of the first winding 17a. Hence, let it be assumed hereinafter that the first winding 17 a ofthe transformer 17 is represented by a series circuit of the leakageinductance LL and the excitation inductance LX1, and that the secondwinding 17 b is represented by the excitation inductance LX2. Therefore,in the current doubler rectifying and smoothing circuit 3, as shown inFIG. 1, the transformer 17 is represented as an equivalent circuit inwhich the series circuit of the leakage inductance LL and the excitationinductance LX1 is connected to opposite ends a and b of the firstwinding 17 a, and the excitation inductance LX2 is connected to oppositeends c and d of the second winding 17 b.

Further, the transformer 17 is constructed by using ferrite cores 22 and22, for instance, which are formed with gaps 21 and 21 and constitute,as a whole, a generally O-shaped ring core, as shown in FIG. 6A. Asshown in FIG. 6B, a conducting wire 23 as the first winding 17 a iswound around one of core portions of the ferrite cores 22 and 22, and aconducting wire 24 as the second winding 17 b is wound around the otherof the core portions of the ferrite cores 22 and 22, in a manner spacedfrom the conducting wire 23 by a predetermined distance. In thistransformer 17, the widths of the gaps 21 and 21 are adjusted, wherebythe excitation inductances LX1 and LX2 can be adjusted in magnitude,while the ferrite cores 22 can be changed in shape to adjust theseparation distance between the conducting wire 23 and conducting wire24, whereby the leakage inductance LL (i.e. the leakage inductance LL1and the leakage inductance LL2) can be adjusted in magnitude. Morespecifically, to increase the magnitude of the leakage inductance LL,the ferrite cores 22 which provides an increased separation distancebetween the conducting wires 23 and 24 is employed, whereas to increasethe magnitude of the excitation inductance LX, the gaps 21 and 21 aremade smaller. Therefore, by adjusting the widths of the gaps 21 and 21and the separation distance between the conducting wires 23 and 24,respectively, the magnitudes of the leakage inductance LL and theexcitation inductances LX1 and LX2 can be defined as desired.

Further, it is also possible to use a leakage transformer shown in FIG.7A in place of the transformer 17. This transformer 25 is constructed byusing a ferrite core 22 a which has intermediate legs 26 and 26 arrangedin a manner opposed to each other with opposed ends thereof kept fromcontact with each other. In this case, the intermediate legs 26 and 26constitute a magnetic flux bypass passage of the invention, and as shownin the figure, a magnetic flux φ1 passes through the bypass passage,thereby increasing the leakage inductance LL of the transformer 25.Further, as shown in FIG. 7B, intermediate portions of a body of aferrite core 22 b may be connected to each other by an intermediate leg28 formed of a magnetic material having a low magnetic permeability. Inthis case as well, the intermediate leg 28 constitutes the magnetic fluxbypass passage of the invention, and as shown in the figure, a magneticflux φ2 passes through the bypass passage, thereby increasing theleakage inductance LL of the transformer 27. Furthermore, it is alsopossible to constitute a whole of the ferrite cores 22 and 22 by usingferrite cores having a low magnetic permeability. In this case as well,the leakage inductance LL can be increased. Further, as shown in FIG. 8,the leakage inductance LL may be increased by connecting a coil 29 of nocore type to the conducting wire 23 of the transformer 17. In this case,the leakage inductance LL can be adjusted with ease. Further, the coil29 may be formed by winding the conducting wire 23.

Next, the operation of the power supply 1 will be described withreference to FIGS. 1 and 5. It should be noted that in the following,description is made by taking the transformer 17 as an example, which isconstituted by setting a turns ratio between the number of turns of thefirst winding 17 a and that of turns of the second winding 17 b to 1:1.In this case, the excitation inductances LX1 and LX2 become equal toeach other, and hence hereinafter they will also be referred to as theexcitation inductances LX, respectively, unless they are discriminatedfrom each other.

First, in this power supply, push-pull FET circuits, not shown,connected respectively to one end 2 a 1 of the primary winding 2 a ofthe transformer 2 and the other end 2 a 2 thereof are driven at 180degrees out of phase with respect to each other, whereby as shown inFIG. 5, a bipolar voltage VS having a voltage value±Vs is inducedbetween the opposite ends of the secondary winding 2 b of thetransformer 2. In this case, in the period T1 during which one of theFET circuits is controlled to an ON state at a duty ratio D of 25%, ahigh voltage is induced on the side of the one end 2 b 1 of thesecondary winding 2 b during the ON time period TON of the FET, and thisinduced voltage causes a current I1 shown in FIG. 1 to flow through acurrent path of the one end 2 b 1 of the secondary winding 2 b, thediode 11, the load 4, the excitation inductance LX2 in the form of thesecond winding 17 b, and the other end 2 b 2 of the secondary winding 2b. In this state, as shown in FIG. 5, a voltage VLX2 having a voltagevalue (VS−V0=(1−D)/D•V0) and directed as shown in FIG. 1 is generatedbetween opposite ends of the excitation inductance LX2, whereby energyis accumulated in the second winding 17 b. At the same time, as shown inthe same figure and FIG. 1, a voltage VLX1 which is inverted in signwith respect to the voltage VLX2 is generated between opposite ends ofthe excitation inductance LX1. Further, during an OFF time period TOFFof the repetition period T1, the energy accumulated in the secondwinding 17 b causes a current I2 to flow in a direction shown in FIG. 1through a current path of one end c of the second winding 17 b, thediode 12, the load 4, and the other end d of the second winding 17 b.Consequently, the voltage VLX2 between opposite ends of the excitationinductance LX2 is caused to assume a voltage value (−V0), and at thesame time, as shown in FIG. 5, a current Ib varying within a range of acurrent variation width ((1−2D/LL)−(1−D)/LX)•V0/f) flows through theexcitation inductance LX2.

Further, in the period T2 during which the other of the FET circuitscontrolled to an ON state at a duty ratio D of 25%, a voltage is inducedon the side of the other end 2 b 2 of the secondary winding 2 b duringan ON time period TON of the FET, and this induced voltage causes acurrent I3 shown in FIG. 1 to flow through a current path of the otherend 2 b 2 of the secondary winding 2 b, the diode 12, the load 4, thefirst winding 17 a as a series circuit of the excitation inductance LX1and the leakage inductance LL, and the one end 2 b 1 of the secondarywinding 2 b. In this state, as shown in FIG. 5, between the oppositeends a and b of the first winding 17 a, there is generated a sum voltage(which has a maximum value (VS−V0=(1−D)/D•V0) and a minimum value (−V0))resulting from addition of the voltage VLX1 (see FIG. 1) between theopposite ends of the excitation inductance LX1 and a voltage VLL (seethe same figure) between the opposite ends of the leakage inductance LL,whereby energy is accumulated in the first winding 17 a. In this case,the voltage VLL between the opposite ends of the leakage inductance LLis obtained by subtracting the voltage VLX1, shown in FIG. 5, betweenthe opposite ends of the excitation inductance LX1 from the voltage(VLX1+VLL), shown in the figure, between the opposite ends of the firstwinding 17 a. As shown in the same figure, this results in the voltageVLL whose maximum value and minimum value are ((1−2D)•V0/D) and (−2V0)respectively, and the period of which is T1/2 (=T2/2).

Further, during an OFF time period TOFF of the period T2, the energyaccumulated in the first winding 17 a causes a current I4 to flow in adirection shown in FIG. 1 through a current path of one end a of thefirst winding 17 a, the diode 11, the load 4, and the other end b of thefirst winding 17 a. Consequently, as shown in FIG. 5, a current Iavarying within a range of a current variation width ((1−2D/LL)•V0/f)flows through the first winding 17 a. It should be noted that as shownin FIGS. 1 and 5, a ripple current IC varying within a range of a verysmall current variation width (2•(1−2D/LL)−(1−D)/LX)•V0/f) flows througha capacitor 13. In the above process of operation, each of averagecurrent values of the currents Ia and Ib becomes equal to one half of anoutput current I0 (see FIG. 18), since a sum total of the current valuesof the currents Ia and Ib becomes equal to the output current I0 and atthe same time the current values thereof are equal to each other. As aresult, magnetic fluxes generated by the current Ia and the current Ibflowing cancel each other. Further, in terms of an instantaneousvariation as well, as shown in FIG. 5, an exciting current IT flowingthrough the transformer 17 varies only slightly within a range of acurrent variation width ((1−D)•V0/(LX•f) resulting from mutualcancellation of the current Ib shown in the figure and the current Iashown in the figure.

As described above, according to the current doubler rectifying andsmoothing circuit 3, the first winding 17 a and the second winding 17 bof the transformer 17 are used as smoothing coils, whereby magneticfluxes generated by respective currents (I0/2) flowing through thewindings 17 a and 17 b cancel each other. Hence, a DC component of theexciting current IT flowing through the transformer 17 becomesapproximately equal to 0 A. Consequently, magnetic saturation in thetransformer 17 caused by a DC bias can be prevented. This makes itpossible to constitute smoothing coils including a large excitationinductance LX by using ferrite cores 22 and 22 having small effectivevolumes, so that the current doubler rectifying and smoothing circuit 3can be caused to serve as more excellent smoothing filters. Further,since the two choke coils 14 and 15 in the conventional current doublerrectifying and smoothing circuit 82 are replaced by one transformer 17,the current doubler rectifying and smoothing circuit 3, and further thepower supply 1 can be made smaller in size.

(Second Embodiment)

Next, the arrangement of a power supply will be described in detail withreference to FIGS. 9 and 10. It should be noted that component partshaving the same functions as those of the power supply 1 are designatedby identical reference numerals, and redundant description thereof isomitted.

A power supply 31 shown in FIG. 9 is an active clamp double-endedforward converter, and has a DC power supply 32, n-channel FETs 33 and34 which are alternately driven, and a capacitor 35, arranged on theside of a primary winding 2 a of a transformer 2.

In the above power supply 31, a switching signal SS1 shown in the figureis input to the gate of the FET 33, and the FET 33 is controlled to anON state during an ON time period TON of the switching signal SS1. Inthis state, as shown in the figure, a current I11 flows through acurrent path of the DC power supply 32, the primary winding 2 a, thedrain and source of the FET 33, and the DC power supply 32, and thecurrent I11 induces, as shown in FIG. 10, a voltage having a voltagevalue (+V0/D) and directed as shown in FIG. 9 between the opposite endsof the secondary winding 2 b. It should be noted that a bipolar voltageof the invention is formed by this voltage and a voltage induced betweenthe opposite ends of the secondary winding 2 b during an ON time periodof a switching signal SS2, referred to hereinbelow. Hereinafter, bothvoltages induced during the period T are generically referred to as abipolar voltage VS. In this embodiment, the bipolar voltage VS generatedbetween the opposite ends of the secondary winding 2 b of thetransformer 2 of the power supply 31 is distinguished from the bipolarvoltage VS generated in the power supply 1 in that voltage waveformsthereof on a plus side and on a minus side are asymmetric.

Next, similarly to the power supply 1, the bipolar voltage VS causes acurrent I1 shown in FIG. 9 to flow, and as shown in FIG. 10, a voltageVLX2 having a voltage value (VS−V0=(1−D)•V0/D) and directed as shown inFIG. 9 is generated between opposite ends of a second winding 17 b,whereby energy is accumulated in the second winding 17 b. At the sametime, as shown in FIG. 10, a voltage VLX1 having a voltage value(−(1−D)•V0/D) which is reversed in sign with respect to the voltage VLX2is generated between the opposite ends of an excitation inductance LX1.Further, during an OFF time period TOFF of the switching signal SS1, theenergy accumulated in the second winding 17 b causes a current I2 toflow in a direction shown in FIG. 9. Consequently, the voltage VLX2 andthe voltage VLX1 are caused to have a voltage (−V0) and a voltage (V0),respectively, and at the same time, as shown in FIG. 10, a current Ibvarying within a range of a current variation width((1−2D/LL)−(1−D)/LX)•V0/f) flows through the second wiring 17 b.

Next, during the OFF time period TOFF of the switching signal SS1, theswitching signal SS2 is an ON time period TON, during which, as shown inFIG. 9, an energy accumulated in the primary winding 2 a causes acurrent I12 to flow through a current path of the primary winding 2 a,the source and drain of the FET 34, the capacitor 35, and the primarywinding 2 a, whereby energy is accumulated in the capacitor 35. On theother hand, when the energy accumulated in the primary winding 2 a isreleased, the energy accumulated in the capacitor 35 causes a currentI13 to flow through a current path of the capacitor 35, the drain andsource of the FET 34, the primary winding 2 a, and the capacitor 35.

During the above ON time period TON of the switching signal SS2, thebipolar voltage VS is induced between the opposite ends of the secondarywinding 2 b, and similarly to the power supply 1, the bipolar voltage VScauses a current I3 shown in FIG. 9 to flow. In this state, as shown inFIG. 10, a sum voltage of the voltages VLX1 and XLV2 directed as shownin FIG. 9 (which has a maximum value (D•V0/(1−D)=VS−V0) and a minimumvalue (−V0)) is generated, whereby energy is accumulated in the firstwinding 17 a. In this case, a voltage VLL between opposite ends of aleakage inductance LL is equal to a value obtained by subtracting thevoltage VLX1, shown in FIG. 10, from the voltage (VLX1+VLL), shown inthe figure. As shown in the figure, this results in the voltage VLLwhose maximum value and minimum value are ((1−2D)•V0/D) and(−(1−2D)•V0/(1−D)), respectively, and the cycle of which is T.

Further, similarly to the power supply 1, during an OFF time period TOFFof the switching signal SS2, the energy accumulated in the first winding17 a causes a current I4 to flow in a direction shown in FIG. 9.Consequently, as shown in FIG. 10, a current Ia varying within a rangeof a current variation width ((1−2D/LL)•V0/f) flows through the firstwinding 17 a. It should be noted that as shown in FIGS. 9 and 10, aripple current IC varying within a range of a very small currentvariation width (2•(1−2D/LL)−(1−D)/LX)•V0/f) flows through a capacitor13.

In the above process of operation, an average current value of each ofthe currents Ia and Ib becomes equal to one half of the output currentI0 shown in FIG. 9, since a sum total of the current values of thecurrents Ia and Ib becomes equal to the output current I0 and at thesame time the current values thereof are equal to each other. As aresult, in the power supply 31 as well, magnetic fluxes generated by thecurrent Ia and the current Ib flowing through the windings 17 a and 17 bcancel each other. Further, in terms of an instantaneous variation aswell, as shown in FIG. 10, an exciting current IT flowing through thetransformer 17 varies only slightly within a range of a currentvariation width ((1−D)•V0/(LX•f)) resulting from mutual cancellation ofthe current Ib appearing in the figure and the current Ia appearing inthe figure.

As described above, in the power supply 31 as well, magnetic saturationin the transformer 17 can be prevented, similarly to the power supply 1.This makes it possible to constitute smoothing coils including the largeexcitation inductance LX by using ferrite cores 22 and 22 having smalleffective volumes, so that it is possible to cause the current doublerrectifying and smoothing circuit 3 to serve as more excellent smoothingfilters, and at the same time the current doubler rectifying andsmoothing circuit 3, and further the power supply 1 can be made smallerin size.

(Third Embodiment)

It should be noted that the present invention is not limited to thearrangement of the above power supply 31, but it is possible to changethe arrangement of the primary winding-side of the transformer 2 asrequired. For instance, as in a power supply 41 shown in FIG. 11, theinvention can be constructed by using a push-pull converter which iscomprised of a transformer 42 having two primary windings 42 a and 42 band a secondary winding 42 c, a FET 43 connected in parallel with aseries circuit of the primary winding 42 a and a DC power source 32, andFET 44 connected in parallel with the series circuit of the primarywinding 42 b and the DC power source 32. In this case, as shown in FIG.12, switching signals SS3 and SS4 for driving the FETs 43 and 44 at 180degrees out of phase with respect to each other are input to the gatesof the FETs 43 and 44 respectively. Further, this power supply 41, andthe current doubler rectifying and smoothing circuit 3 for use withvarious kinds of power supplies described hereinafter operate in thesame manner as the current doubler rectifying and smoothing circuit 3 ofthe power supply 1. Hence, the same component parts as those of thepower supply 1 are designated by identical reference numerals, anddescription of operations as power supplies is omitted.

(Fourth Embodiment)

Further, as in a power supply 51 shown in FIG. 13, the invention canalso be constructed by using a so-called half-bridge converter in whichon the side of a primary winding 2 a, a series circuit of two FETs 43and 44 and a series circuit of two capacitors 52 and 53 are connected inparallel with a DC power source 32, while opposite ends 2 a 2 and 2 a 1of the primary winding 2 a are connected to a junction of the FETs 43and 44 and a junction of the capacitors 52 and 53, respectively. In thiscase as well, the switching signals SS3 and SS4 appearing in FIG. 12 areinput to the gates of the FETs 43 and 44 respectively.

(Fifth Embodiment)

Further, as in a power supply 61 shown in FIG. 14, the invention canalso be constructed by using a so-called full-bridge converter, in whichon the side of a primary winding 2 a, a series circuit of two FETs 62and 63 and a series circuits of two FETs 64 and 65 are connected inparallel with a DC power source 32, while opposite ends 2 a 1 and 2 a 2of the primary winding 2 a are connected to a junction of the FETs 62and 63 and a junction of the FETs 64 and 65, respectively. In this case,the switching signal SS3 appearing in FIG. 12 is input to the gates ofthe FETs 62 and 65, and the switching signal SS4 appearing in FIG. 12 isinput to the gates of the FETs 63 and 64.

(Sixth Embodiment)

Further, as in a power supply 71 shown in FIG. 15, the invention canalso be constructed by using a so-called asymmetric half-bridgeconverter in which on the side of a primary winding 2 a, a seriescircuit of two FETs 72 and 73 is connected in parallel with a DC powersource 32, while a series circuit of a capacitor 74 and the primarywinding 2 a is connected in parallel between the drain and the source ofthe FET 73. In this case, the switching signals SS1 and SS2 appearing inFIG. 9 are input to the gates of the FETs 72 and 73, respectively.

Further, although in each of the above power supplies, the examples inwhich the FETs are used as the switching elements arranged on the sideof the primary winding of the switching transformer were described, thisis not limitative, but it is possible to employ various types ofswitching elements, such as transistors and the like. Further, althoughin the embodiments of the invention, description was made based on theexamples in which the transformer 17 is constituted by setting a turnsratio between the number of turns of the first winding 17 a and that ofturns of the second winding 17 b to 1:1, this is not limitative, but thetransformer used can be constituted at an arbitrary turns ratio. In thiscase, the amount of the ripple current IC flowing through the capacitor13 can be determined as desired by selecting a turns ratio as required.

INDUSTRIAL APPLICABILITY

As described above, according to the rectifying and smoothing circuit ofthe invention, the first winding and the second winding of the secondtransformer, which are wound in a manner permitting magnetic fluxesgenerated by currents flowing therethrough to cancel each other, areused as the first and second smoothing inductors, whereby an excitingcurrent flowing through the second transformer can be sharply decreased,and at the same time magnetic saturation in the second transformer canbe prevented. This makes it possible to construct a smoothing inductorhaving a larger excitation inductance by using cores having smalleffective volumes. Therefore, the rectifying and smoothing circuit canserve as more excellent smoothing filters, and at the same time themanufacturing costs and size thereof can be reduced through reduction ofthe number of component parts thereof. As a result, a power supplyreduced in manufacturing costs and size can be realized by using therectifying and smoothing circuit.

Further, according to the double-ended converter of the invention, it ispossible to realize a power supply having more excellent smoothingfilters and reduced in manufacturing costs and size by reduction of thenumber of component parts of the rectifying and smoothing circuit.

What is claimed is:
 1. A rectifying and smoothing circuit comprising: afirst switching transformer having an output winding having one end andanother end; a low potential-side output portion; a high potential-sideoutput portion; a first inductor connected between said one end of saidoutput winding and said low potential-side output portion; a secondinductor connected between said another end of said output winding andsaid low potential-side output portion; a first rectifying elementconnected between said one end of said output winding and said highpotential-side output portion; and a second rectifying element connectedbetween said another end of said output winding and said highpotential-side output portion, the rectifying and smoothing circuitgenerating a DC voltage by rectifying and smoothing a bipolar voltageinduced across said output winding, wherein said first inductor and saidsecond inductor are constructed by a first winding and a second windingof a second transformer, respectively, said first winding and saidsecond winding being wound in a manner such that magnetic fluxesgenerated by respective currents flowing therethrough cancel each other.2. The rectifying and smoothing circuit according to claim 1, whereinsaid first inductor and said second inductor are each constructed of aseries circuit of an equivalent leakage inductance of said secondtransformer and an equivalent excitation inductance of said secondtransformer.
 3. The rectifying and smoothing circuit according to claim2, wherein said first winding and said second winding of said secondtransformer are wound so as to be spaced from each other by apredetermined distance.
 4. The rectifying and smoothing circuitaccording to claim 2, wherein said second transformer includes amagnetic flux bypass passage.
 5. The rectifying and smoothing circuitaccording to claim 2, wherein said second transformer uses magneticcores having a low magnetic permeability.
 6. The rectifying andsmoothing circuit according to claim 2, wherein said first winding andsaid second winding of said second transformer are wound around magneticcores formed with gaps.
 7. The rectifying and smoothing circuitaccording to claim 2, including at least one third inductor eachconnected in series with a corresponding one of said first winding andsaid second winding.
 8. The rectifying and smoothing circuit accordingto claim 1, wherein said first winding and said second winding of saidsecond transformer are wound so as to be spaced from each other by apredetermined distance.
 9. The rectifying and smoothing circuitaccording to claim 1, wherein said second transformer includes amagnetic flux bypass passage.
 10. The rectifying and smoothing circuitaccording to claim 1, wherein said second transformer uses magneticcores having a low magnetic permeability.
 11. The rectifying andsmoothing circuit according to claim 1, wherein said first winding andsaid second winding of said second transformer are wound around magneticcores formed with gaps.
 12. The rectifying and smoothing circuitaccording to claim 1, including at least one third inductor eachconnected in series with a corresponding one of said first winding andsaid second winding.
 13. A double-ended converter comprising arectifying and smoothing circuit, said rectifying and smoothing circuitincluding: a first switching transformer having an output winding havingone end and another end; a low potential-side output portion; a highpotential-side output portion; a first inductor connected between saidone end of said output winding and said low potential-side outputportion; a second inductor connected between said another end of saidoutput winding and said low potential-side output portion; a firstrectifying element connected between said one end of said output windingand said high potential-side output portion; and a second rectifyingelement connected between said another end of said output winding andsaid high potential-side output portion, said rectifying and smoothingcircuit generating a DC voltage by rectifying and smoothing a bipolarvoltage induced across said output winding, wherein said first inductorand said second inductor are constructed by a first winding and a secondwinding of a second transformer, respectively, said first winding andsaid second winding being wound in a manner such that magnetic fluxesgenerated by respective currents flowing therethrough cancel each other.14. The double-ended converter according to claim 13, wherein said firstinductor and said second inductor are each constructed by a seriescircuit of an equivalent leakage inductance of said second transformerand an equivalent excitation inductance thereof.